Oscillator with primary and secondary lc circuits

ABSTRACT

One aspect of this disclosure is an apparatus including an oscillator that includes a secondary LC circuit to increase a tuning range of the oscillator and/or to reduce a phase noise of the oscillator. Another aspect of this disclosure is an apparatus that includes oscillator with a primary LC circuit and a secondary LC circuit. This oscillator can operate in a primary oscillation mode or a secondary oscillation mode, depending on whether oscillation is set by the primary LC circuit or the secondary LC circuit.

CROSS REFERENCE TO RELATED APPLICATION

This application is a non-provisional of and claims the benefit under 35U.S.C. §119(e) of U.S. Provisional App. No. 61/779,386, filed Mar. 13,2013, titled “OSCILLATOR WITH PRIMARY AND SECONDARY LC CIRCUITS,” theentire disclosure of which is hereby incorporated by reference herein.

TECHNICAL FIELD

The invention relates to electronics, and, more particularly, tocircuits configured to oscillate.

DESCRIPTION OF THE RELATED TECHNOLOGY

Electronic systems, such as transceivers that include a synthesizer, caninclude an oscillator. In some applications, an oscillator can be usedfor modulation and/or demodulation in a transceiver. In otherapplications, an oscillator can be used for a clock and data recoverycircuit and/or in a phase-locked loop. A number of oscillators cangenerate a relatively high frequency output signal, which can bedesirable for various wired and wireless applications. An LCresonator-based oscillator is one type of oscillator that can generate arelatively high frequency output.

Noise from an oscillator can impact the performance of an electronicsystem, such as a transceiver. One type of noise in an oscillator isphase noise. Phase noise can be a frequency domain representation ofshort-term fluctuations in a phase of a waveform caused by time domaininstabilities. Phase noise can represent a power spectral density of aphase of a signal and/or the power spectral density of the signal.Numerous attempts have been made to reduce the phase noise ofoscillators, such as LC oscillators. However, noise issues persist inoscillators. For example, some transceivers and/or components oftransceivers have stringent noise requirements that can be difficult tomeet due to phase noise generated by LC oscillators. Because noiseissues can impact the performance of electronic systems, better noiseperformance of a part can contribute to commercial success of the partand/or electronic system. Accordingly, there is a need for low noiseoscillators.

SUMMARY

One aspect of this disclosure is an apparatus that includes anoscillator. The oscillator includes a primary LC circuit, a sustainingamplifier, and a secondary LC circuit. The primary LC circuit has afirst node and a second node. The sustaining amplifier comprises a firsttransistor configured to drive the first node of the primary LC circuitin response to an input at a first control terminal of the firsttransistor. The secondary LC circuit comprises a tuning network havingan adjustable impedance. The tuning network is electrically coupled tothe first control terminal of the first transistor. The secondary LCcircuit is configured to pass a bias to the first control terminal ofthe first transistor.

In certain embodiments, the oscillator is configured to causeoscillation of a signal at the first node to be set by the secondary LCcircuit by adjusting the adjustable impedance. According to some ofthese embodiments, the primary LC circuit comprises a switching networkconfigured to selectively adjust an impedance, such as a capacitance,between the first node and the second node, and the oscillator isconfigured to cause the primary LC circuit to set oscillation of thesignal at the first node by adjusting the adjustable impedance of thesecondary LC circuit and adjusting the impedance between the first nodeand the second node with the switching network. The apparatus caninclude an impedance element, such as a capacitor, in parallel with theswitching network.

During operation, a signal at the control terminal of the firsttransistor can have a greater amplitude than the signal at the firstnode.

The tuning network can adjust a capacitance of the secondary LC circuitand/or adjust an inductance of the secondary LC circuit. The tuningnetwork can be configured to tune a resonant frequency of the secondaryLC circuit.

The secondary LC circuit can cause a tuning range of the oscillator tobe extended. The secondary LC circuit can pass the bias to the firstcontrol terminal of the first transistor via an inductor. The biaspassed by the secondary LC circuit can be at a ground potential or apower rail potential.

The sustaining amplifier can further comprise a second transistorconfigured to drive the second node of the primary LC circuit inresponse to an input at a second control terminal of the secondtransistor, and the secondary LC circuit can be coupled to the secondcontrol terminal of the second transistor. The tuning network can beelectrically coupled between the first control terminal of the firsttransistor and the second control terminal of the second transistor, andthe tuning network can be configured to adjust the impedance between thefirst control terminal of the first transistor and the second controlterminal of the second transistor. The secondary LC circuit can cause anamount of time that the first transistor drives the first node while thesecond transistor drives the second node to be reduced.

The oscillator can be a voltage-controlled oscillator.

Another aspect of this disclosure is an apparatus that includes anoscillator. The oscillator includes a primary LC circuit having a firstnode and a second node. The oscillator also includes a differential pairof field effect transistors. The differential pair of field effecttransistors includes a first field effect transistor having a firstdrain coupled to the first node and a first gate coupled to the secondnode via a first capacitor and a second field effect transistor having asecond drain coupled to the second node and a second gate coupled to thefirst node via a second capacitor. The oscillator further includes asecondary LC circuit coupled to the first gate of the first field effecttransistor and the second gate of the second field effect transistor.The oscillator is configured to generate a first signal at the firstgate having greater amplitude than a second signal at the first drain.

In certain embodiments, the secondary LC circuit comprises a tunablepassive impedance network configured to adjust an impedance between thefirst gate of the first field effect transistor and the second fieldeffect transistor. According to some of these embodiments, the primaryLC circuit comprises a switching network configured to selectivelyelectrically couple capacitance between the first node and the secondnode.

Yet another aspect of this disclosure is an apparatus that includes anoscillator. The oscillator includes a sustaining amplifier; a first LCcircuit coupled to the sustaining amplifier; the sustaining amplifierconfigured to sustain oscillation of the first LC circuit; and a secondLC circuit coupled to the sustaining amplifier. The oscillator isconfigured to operate in at least a first oscillation mode and a secondoscillation mode, in which oscillation of the oscillator is set by thefirst LC circuit in the first oscillation mode, and in which oscillationof the oscillator is set by the second LC circuit is the secondoscillation mode.

In certain embodiments, the sustaining amplifier comprises a firsttransistor configured to drive a node of the first LC circuit inresponse to first signal received at a control terminal of the firsttransistor, and the second LC circuit is coupled to the control terminalof the first transistor.

According to various embodiments, the sustaining amplifier comprises afirst field effect transistor having a first drain and a first gate anda second field effect transistor having a second drain and a secondgate; wherein the primary LC circuit has a first node and a second node,wherein the first drain is coupled to the first node, wherein the firstgate is coupled to the second node via a first capacitor, wherein thesecond drain is coupled to the second node, wherein the second gate iscoupled to the first node via a second capacitor, and wherein the secondLC circuit is coupled to the first gate and the second gate.

For purposes of summarizing the disclosure, certain aspects, advantagesand novel features of the inventions have been described herein. It isto be understood that not necessarily all such advantages may beachieved in accordance with any particular embodiment of the invention.Thus, the invention may be embodied or carried out in a manner thatachieves or optimizes one advantage or group of advantages as taughtherein without necessarily achieving other advantages as may be taughtor suggested herein.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating an oscillator.

FIG. 2 is graph illustrating thermal noise current in a sustainingamplifier of the oscillator of FIG. 1.

FIG. 3A is a schematic diagram illustrating a passive impedance networkconfigured to bias control inputs of switches in a sustaining amplifier,according to an embodiment.

FIG. 3B is a schematic diagram illustrating another passive impedancenetwork configured to bias control inputs of switches in a sustainingamplifier, according to another embodiment.

FIG. 3C is a schematic diagram illustrating another passive impedancenetwork configured to bias control an input of a switch in a singlesided sustaining amplifier, according to another embodiment.

FIG. 4 is a schematic diagram of an oscillator including a passiveimpedance network configured to bias a sustaining amplifier, accordingto an embodiment.

FIG. 5 shows graphs illustrating a relationship among transconductancesand current at the drains of differential transistors in a sustainingamplifier that indicate a reduction in zero crossing noise according toan embodiment.

FIGS. 6A and 6B are graphs illustrating relationships among phase noiseand frequency that show improved noise performance for oscillators withpassive impedance networks configured to bias sustaining amplifiers.

FIG. 7 is a block diagram of a switching network.

FIG. 8A is a schematic diagram of a resonant circuit including oneswitching circuit according to an embodiment.

FIG. 8B is a schematic diagram of a resonant circuit including oneswitching circuit according to another embodiment.

FIG. 9 is a schematic diagram of a switching circuit according to anembodiment.

FIGS. 10A and 10B are graphs illustrating relationships among noisevoltage spectral density in switching circuits showing a reduction innoise generated by a switching circuit according to an embodiment.

FIGS. 11A and 11B are graphs illustrating voltage swings in switchingcircuits showing that voltage swings stay within a desired range ofbreakdown voltages according to an embodiment.

FIG. 12 is a schematic diagram of an oscillator having a secondary LCtuning network, according to an embodiment.

FIGS. 13A and 13B are schematic diagrams of an oscillator withdual-oscillation modes, according to an embodiment.

FIG. 14 is a schematic diagram illustrating a tunable passive impedancenetwork configured to bias control an input of a switch in a singlesided sustaining amplifier, according to an embodiment.

DETAILED DESCRIPTION OF CERTAIN EMBODIMENTS

The following detailed description of certain embodiments presentsvarious descriptions of specific embodiments of the inventions. However,the inventions can be embodied in a multitude of different ways asdefined and covered by the claims. In this description, reference ismade to the drawings where like reference numerals indicate identical orfunctionally similar elements.

Generally described, aspects of this disclosure relate to low noiseoscillators. More specifically, aspects of this disclosure relate tooscillators with low noise in a sustaining amplifier and/or a switchingcircuit configured to adjust a resonant frequency of the oscillator.Thermal noise current can be reduced in such oscillators, which canresult in less phase noise.

It can be desirable for an oscillator to have relatively good noiseperformance. Noise can impact a number of performance aspects of acommunications transceiver or other electronic system. For instance,noise from an oscillator can impact receiver blocking performance in awireless communications transceiver and/or a transmitter spectral mask.As another example, oscillator noise can also affect jitter performanceof a clock and data recovery circuit. Accordingly, there is a need foroscillators with relatively low noise in a wide variety of applications.

A number of emerging wired and wireless applications benefit fromoperating at relatively high frequencies, such as frequencies in theradio frequency (RF) range. One type of oscillator suitable for suchapplications is an LC resonator-based oscillator. The apparatus,systems, and methods described herein relate to reducing noise, such asphase noise, of LC resonator-based oscillators and/or noise in otheroscillators.

FIG. 1 is a block diagram illustrating an oscillator 10. The oscillator10 can include one or more of a resonant circuit 12 that can include aswitching network 14, a sustaining amplifier 16, and a bias source 18.The oscillator 10 can generate a periodic electronic signal. Such anoutput of the oscillator 10 can be used in any application with a needfor a periodic electronic signal, such as modulating/demodulating asignal and/or a clock and data circuit. In some implementations, theoscillator 10 can be configured to generate a sinusoidal output signal.The oscillator 10 can be a voltage-controlled oscillator (VCO),according to some implementations.

The resonant circuit 12 can be any circuit configured to oscillate togenerate a periodic signal. Although some of the description hereinrelates to an LC tank for illustrative purposes, it will be understoodthat the principles and advantages described herein can be applied to anumber of other resonant circuits including, for example, RCoscillators, Colpitts oscillators, Armstrong oscillators, Pierceoscillators, Clapp oscillators, relaxation oscillators, the like, or anycombination thereof.

The resonant circuit 12 can generate a range of output frequencies. Morespecifically, the switching network 14 can obtain one or more controlsignals to adjust the output frequency of the oscillator 10. Based onthe one or more control signals, the resonant circuit 12 can oscillateat a higher or a lower frequency. In an LC tank implementation of theresonant circuit 12, a resonant frequency ω of the resonant circuit 12can be proportional to the reciprocal of the square root of theinductance L times the capacitance C, for example, as represented by theEquation 1.

$\begin{matrix}{\omega = \frac{1}{\sqrt{L\; C}}} & \left( {{Eq}.\mspace{14mu} 1} \right)\end{matrix}$

The resonant circuit 12 can generate periodic signals at a first node N1and a second node N2. For example, the voltage at the first node N1 andthe second node n2 can be periodic as the resonant circuit 12oscillates. The signals at the first node N1 and the second node N2 canbe sinusoidal signals that are 180 degrees out of phase with each other,in some implementations. For instance, the first node N1 and the secondnode N2 can have voltages that have opposite signs and approximately thesame magnitude at any given time. In other implementations, the firstnode N1 and the second node N2 can have voltages that have oppositelogical values at any given time. In some implementations, the firstnode N1 and the second node N2 can be referred to as a non-inverted nodeand an inverted node, respectively, the signals can have values that areinverted from each other.

The resonant circuit 12 can include the switching network 14. Theswitching network 14 can include one or more circuit elements that canbe coupled in parallel and/or series with a resonant portion of theresonant circuit 12 configured to oscillate. This can adjust theresonant frequency ω of the oscillator 10. For example, in an LC tankimplementation, the switching network 14 can include one or morecapacitive circuit elements that can be coupled in parallel with othercapacitive elements of the LC tank via switches, such as field effecttransistors. Based on one or more control signals provided to theswitches, more or less capacitance can be coupled in parallel with theLC tank. With additional capacitance, the output frequency can decrease.Conversely, with reduced capacitance, the output frequency can increase.Coupling one or more circuit elements across the portion of the resonantcircuit 12 configured to oscillate can cause the resonant circuit 12 tooscillate within a selected frequency band.

The sustaining amplifier 16 can compensate for energy losses and/ordissipation in the resonant circuit 12, thereby keeping the resonantcircuit 12 oscillating at the resonant frequency ω. For instance, in anLC tank implementation, the sustaining amplifier 16 can include a firsttransistor configured to drive a first node N1 of the LC tank based on avoltage on a second node N2 of the LC tank and a second transistorconfigured to drive the second node N2 of the LC tank based on a voltageon the first node N1 of the LC tank.

The sustaining amplifier 16 can be biased by any suitable bias source18. For instance, the bias source 18 can be a bias current source. Thebias source 18 can provide a bias current which can be passed bytransistors in the sustaining amplifier 16 to the resonant circuit 12.In some implementations, the bias source 18 can include a transistorconfigured to pass a voltage from a power rail (e.g., ground for aN-type device and power for a P-type device) based on a bias voltageapplied to the transistor, for example, at the gate of a field effecttransistor.

Numerous attempts have been made to reduce the phase noise ofoscillators, and that of LC oscillators in particular. However, at leasttwo noise sources appear to be unaddressed. These two noise sourcesinclude a first noise source due to the active devices of a resonantsustaining amplifier and a second noise source due to a capacitorswitching network used to tune a resonant frequency of LC resonantcircuit. These two noise sources can be of particular importance in a1/f² region around an offset frequency from the carrier f_(o)/f², inwhich f can represent an offset from resonant frequency of an LCoscillator, such as an LC tank. For example, f_(o) can be approximately

$2\pi*\frac{1}{\sqrt{L\; C}}$

for an LC tank, in which L can represent an inductance of the LC tankand C can represent a capacitance of the LC tank. On a log plot of phasenoise versus offset frequency from the carrier, the 1/f² region can havea fixed slope of −20 dB/dec for LC oscillators. The 1/f² region cancorrespond to thermal noise of an oscillator. Accordingly, reducingthermal noise of the oscillator can reduce noise in the 1/f² region. Oneor more aspects of the systems, apparatus, and methods provided hereincan, among other things, reduce noise generated by the first noisesource and/or to reduce noise generated by the second noise source.

Although non-silicon based processes, such as GaAs processes, have beenused for some conventional oscillators with low phase noise, providinglow phase noise and/or ultra-low phase noise oscillators on a CMOSand/or a BiCMOS process is desirable. One advantage of implementing anoscillator in CMOS and/or SiCMOS is that such an oscillator can beintegrated along with other circuitry formed by the CMOS and/or SiCMOSprocess. For example, it is projected that a need will exist in thecontext of base stations to integrate several components onto a singlechip, such as one or more analog-to-digital converters (ADCs), one ormore frequency synthesizers each having at least one voltage-controlledoscillator (VCO), one or more digital signal processors (DSPs), thelike, or any combination thereof. A CMOS and/or BiCMOS process canprovide a cost effective way to implement such integration. Reducingphase noise of an oscillator such that the oscillator can meet noisespecifications and be manufactured via a CMOS and/or BiCMOS process isone object, among others, of one or more aspects of the technologydescribed in this disclosure. However, it will be understood that thecircuits illustrated and/or described herein can be manufactured via anysuitable process.

Phase noise from the first noise source described earlier can be reducedvia a passive impedance network, which can bias active devices of thesustaining amplifier such that a conduction angle of the sustainingamplifier is decreased. For instance, a differential inductor can beconfigured to implement such biasing. This biasing can reduce an amountof radio frequency (RF) current and/or a duration of time for which theRF current is injected to or from the resonant circuit 12. The passiveimpedance network can include an inductor, which can create a secondresonant circuit with capacitance of a sustaining amplifier. The secondresonant circuit can be coupled to the gate and/or drain of a fieldeffect transistor of the sustaining amplifier, in some implementations.In the second resonant circuit, the inductor can resonate a capacitanceassociated with the sustaining amplifier. For example, the inductor canresonate the capacitance associated with the sustaining amplifier so asto increase the tenability of the resonant circuit 12. As anotherexample, the inductor can resonate the capacitance associated with thesustaining amplifier so as to reduce phase noise of the oscillator. Asyet another example, the inductor can resonate the capacitanceassociated with the sustaining amplifier so as to reduce a conductionangle of the sustaining amplifier. The capacitance associated with thesustaining amplifier can be a parasitic capacitance of the sustainingamplifier. More detail regarding reducing phase noise from the firstnoise source will be provided below, for example, with reference toFIGS. 3A-6B.

Phase noise from the second source described earlier can be reduced viaa switching circuit (for example, a capacitor switching circuit) thatincludes active circuit elements, such as transistors, configured toreduce the contribution of switch noise to the phase noise of theoscillator. Reducing phase noise generated by the second source appearsto have been ignored and/or not addressed in the relevant literature.More detail regarding reducing phase noise from the second noise sourcewill be provided below, for example, with reference to FIGS. 8A-11D. Anycombination of features described herein with reference to an oscillatorconfigured to reduce noise from the first noise source can beimplemented in conjunction with an oscillator having any combination offeatures described herein related to reducing noise from the secondnoise source.

The headings provided herein are provided for convenience only and donot necessarily affect the scope or meaning of the claimed invention.

Reducing Noise Generated by Sustaining Amplifier

Thermal noise current of a sustaining amplifier of an oscillator cancause and/or contribute to the first noise source described earlier. Thethermal noise current can be represented by an RF current waveform ofthe oscillator, for example, as shown in FIG. 2.

FIG. 2 is graph illustrating thermal noise current in a sustainingamplifier of the oscillator of FIG. 1. Transistors Mp and Mn canrepresent opposing transistors in a differential pair of transistors ofthe sustaining amplifier, for example, as shown in the embodimentsillustrated in any of FIGS. 3A, 3B, 3C and/or 4. The darker line in theMp graph represents current at the drain of Mp while Mp is on, and thelighter line in the Mp graph represents current at the drain of Mn whileMp is off. Similarly, the darker line in the Mn graph represents currentat the drain of Mn while Mn is on, and the lighter line in the Mn graphrepresents current at the drain of Mp while Mn is off.

FIG. 2 also illustrates the thermal noise current generated bytransistors Mp and Mn while these transistors are on. Such thermal noisecan be injected into a resonant circuit at zero crossing points of anoutput voltage waveform of the oscillator. The thermal noise generatedby the transistor Mp can be injected into a resonant circuit, such as anLC tank, at an inverted node (for example, node N2), and the thermalnoise generated by the transistor Mn can be injected into the resonantcircuit at a non-inverted node (for example, node N1) of the oscillator.The thermal noise current can represent an RF current. The sum of thethermal noise generated by the transistors Mp and Mn can be injectedinto the resonant circuit.

An oscillator can be sensitive to noise injected into an LC tank orother resonant circuit at zero crossing points, in which an outputvoltage waveform of the oscillator crosses zero volts. Noise at zerocrossing points can cause an irrecoverable phase disturbance. A zerocrossing point can occur at a transition between the off state and theon state of the transistors Mp and Mn of the sustaining amplifier. Forexample, a zero crossing point can be represented by the differentialdrain voltages of transistors Mp and Mn of the sustaining amplifier 16both approaching zero volts. Because a transition from the on state tothe off state of transistor Mp or Mn cannot be instantaneous, there is afinite amount of time in which the thermal noise current generated byboth transistors Mp and Mn is present at the sensitive zero crossingpoints. As a result, thermal noise can have the greatest effect on phasenoise when the transistors in the sustaining amplifier are switching onand/or off. A reduction in an amount of time that thermal noise currentis present at the zero crossing instants can result in a reduction in anamount of cyclo-stationary noise injected into the LC tank or otherresonant circuit. Consequently, the overall phase noise of theoscillator can be decreased by reducing noise at zero crossing points.

Another challenge in reducing noise at zero crossing points relates tothe effect of parasitics of the sustaining amplifier on the resonantcircuit. For example, increasing the size of transistors in thesustaining amplifier can increase parasitic capacitance on the LC tank.Increasing fixed capacitance in the LC tank, can reduce the impact ofsimilarly sized tuning capacitors configured to be switched in/out withcapacitor(s) in the LC tank to tune the resonant frequency. To maintaina similar level of tenability, the size of tuning capacitors can beincreased to account for the parasitic capacitance of transistors of thesustaining amplifier. This can increase the phase noise of theoscillator, which can be undesirable.

Phase noise from the first source can be reduced via a passive impedancenetwork, which can include, for example, a differential inductor, tobias active devices of the sustaining amplifier such that a conductionangle of the amplifier can be reduced. The conduction angle, which canalso be referred to as an angle of flow, of an amplifier can represent aportion of the oscillator cycle during which devices of the amplifierconduct current. Alternatively or additionally, an amount of radiofrequency (RF) thermal noise current and/or a duration of time for whichthe RF thermal noise current is injected to or from the LC resonator canbe substantially reduced.

FIG. 3A is a schematic diagram illustrating a passive impedance network20 a configured to bias control inputs of switches in a sustainingamplifier, according to an embodiment. The passive impedance network 20a can apply a bias to control terminals of the switches, such astransistors Mn and Mp, to enable the switches to switch faster at zerocrossing points. For instance, the bias can increase the bias voltageapplied to the control terminals of transistors Mn and Mp. Increasingthe switching speed of the transistors Mn and Mp via such biasing canreduce an amount of time for which both of the switches Mn and Mp are atleast partially on. As a result, at zero crossing points, an amount oftime during which both switches Mn and Mp inject thermal noise into theresonant circuit 12 can be reduced, thereby reducing phase noise in theoscillator 10 a.

The transistor Mn can be configured to drive a non-inverted node (forexample, node N1) of the resonant circuit 12 based on a voltage at aninverted node (for example, node N2) of the resonant circuit 12, and thetransistor Mp can be configured to drive the inverted node of theresonant circuit 12 based on a voltage at the non-inverted node of theresonant circuit 12. When the transistor Mn is an NMOS transistor, thenon-inverted node can be pulled down when the inverted node is high.Similarly, when the transistor Mp is an NMOS transistor, the invertednode can be pulled down when the non-inverted node is high. Although thetransistors Mn and Mp are illustrated as field effect transistors forillustrative purposes, the transistors can be bipolar transistors or anyother suitable transistors, which can be formed by any suitable process.

To increase the switching speed of the transistors Mn and Mp, a gatebias voltage Vg can be applied to their respective gates while the gatebias voltage Vg is electrically isolated from the inverted node and thenon-inverted node of the resonant circuit 12 by the passive impedancenetwork 20 a. Thus, the voltage applied to the gate of transistor Mn canbe based on a voltage at the inverted node and the gate bias voltage Vgand the voltage applied to the gate of the transistor Mp can be based ona voltage at the non-inverted node and the gate bias voltage Vg. Thebias voltage Vg can be generated by an independent bias circuit. In someimplementations, the bias circuit can be programmable and configured tominimize and/or reduce phase noise. The bias applied by the passiveimpedance network 20 a can have relatively low noise. The gate biasvoltage Vg can increase the voltage applied to the gates of thetransistors Mn and Mp beyond the voltage provided from the non-invertednode and the inverted node of the resonant circuit 12 when thetransistors Mn and Mp are p-type transistors. Conversely, when thetransistors Mn and Mp are n-type transistors, the gate bias voltage Vgcan decrease the voltage applied to the gates of the transistors Mn andMp beyond the voltage provided from the non-inverted node and theinverted node of the resonant circuit 12.

The passive impedance network 20 a can include one or more passiveimpedance elements, such as passive impedance elements 32 a, 32 b, and32 c. The passive impedance elements 32 a, 32 b, and 32 c can beexplicit passive impedance elements rather than merely parasiticimpedances. The first explicit passive impedance element 32 a can blocka direct current (DC) voltage from being applied to the gate oftransistor Mp. Similarly, the third explicit passive impedance element32 c can block a DC voltage from being applied to the gate of transistorMn. The first and third explicit passive impedance elements 32 a, 32 ccan include inductors, capacitors, resistors, or other passive circuitelements. In some implementations, the first and third explicit passiveimpedance elements 32 a, 32 c can both be capacitors.

The second explicit passive impedance element 32 b can be configured toprovide a low-noise, high impedance characteristic at the resonantfrequency ω of the resonant circuit 12. In some implementations, theresonant frequency ω can be in the RF frequency range. The low-noise,high impedance characteristic can enable faster switching of thetransistors Mn and Mp. The amount of time that the transistors Mn and/orMp operate in the Ohmic region can also be reduced by applying the gatebias voltage Vg via the second explicit passive impedance element 32 b.The second passive impedance element 32 b can be a differential inductorin some implementations.

In addition to reducing phase noise of the oscillator 10, the passiveimpedance network 20 a can increase the voltage swing at an output ofthe oscillator 10, for example, by reducing common mode current. Thesecond explicit passive impedance element 32 b can reduce parasitics ofthe transistors Mn and Mp on the non-inverted node and the inverted nodeof the resonant circuit 12. This can enable the transistors Mn and Mp tobe relatively large, with relatively large parasitic capacitance inrelation to an effective capacitance of an LC tank, without having asignificant impact on the tenability of the LC tank.

FIG. 3B is a schematic diagram illustrating a passive impedance network20 b configured to bias control inputs of switches in a sustainingamplifier, according to another embodiment. The oscillator 10 billustrated in FIG. 3B has a different passive impedance network fromthe oscillator 10 a in FIG. 3A, otherwise these oscillators aresubstantially the same and/or functionally similar. In the passiveimpedance network 20 b, separate second and fourth explicit passiveimpedance elements 32 b and 32 d can separately apply a bias to gates oftransistors Mn and Mp, respectively. As illustrated in FIG. 3B, thegates of the transistors Mn and Mp can be biased with bias voltagesV_(G2) and V_(G1), respectively. The bias voltages V_(G2) and V_(G1) canhave different voltages in some implementations. In otherimplementations, the gates of transistors Mn and Mp can be biased bybias voltages V_(G2) and V_(G1) having substantially the same voltage.The passive impedance network 20 b can be functionally similar to thepassive impedance network 20 a.

FIG. 3C is a schematic diagram illustrating a passive impedance network20 c configured to bias control an input of a switch in a sustainingamplifier, according to another embodiment. The oscillator 10 cillustrated in FIG. 3C has a single ended sustaining amplifier, insteadof a differential sustaining amplifier like in the oscillators 10 a and10 b. Besides having a single ended sustaining amplifier and a differentpassive impedance network, the oscillator 10 c can be substantially thesame and/or functionally similar to the oscillators 10 a and/or 10 b.

The passive impedance network 20 c can include the second explicitpassive impedance element 32 b and the third explicit passive impedanceelement 32 c, which can include any combination of features describedherein with reference to these explicit passive impedance elements. Forexample, the third explicit passive impedance element 32 c can block adirect current (DC) voltage from being applied to the gate of transistorMn. A first end of the third explicit passive impedance element 32 c canbe coupled to an inverted node (for example, node N2) and a second endof the third explicit passive impedance element 32 c can be coupled to acontrol terminal (such as a gate when Mn in a field effect transistor)of the transistor Mn. The third explicit passive impedance element 32 ccan be a capacitor according to some implementations. In the passiveimpedance network 20 c, the second explicit passive impedance element 32b can apply a bias V_(G) to the gate of the transistor Mn. In someimplementations, the second explicit passive impedance element 32 b canbe an inductor. A first end of the second explicit passive impedanceelement can receive the bias V_(G) and a second end of the secondexplicit passive impedance element 32 b can apply the bias V_(G) to thegate of the transistor Mn. The passive impedance network 20 c can befunctionally similar to the passive impedance networks 20 a and/or 20 b,but in the context of a single ended sustaining amplifier.

FIG. 4 is a schematic diagram of an oscillator 10 including a passiveimpedance network 20 c configured to bias a sustaining amplifier,according to an embodiment. The oscillator 10 illustrated in FIG. 4 isin an LC tank configuration. The LC tank can include first and secondinductors 41 a and 41 b and a capacitor switching network 14.

The inductive circuit elements of the LC tank can include the first andsecond inductors 41 a and 41 b. The first inductor 41 a can include afirst end coupled to a power rail (for example, ground) and a second endcoupled to the first node N1. The second inductor 41 b can include afirst end coupled to a power rail (for example, ground) and a second endcoupled to the second node N2. The effective inductance of the LC tankcan be based on inductance values of the first inductor 41 a and thesecond inductor 41 b.

The capacitor switching network 14 can adjust the resonant frequency ofthe oscillator 10. The capacitor switching network 14 can include aplurality of capacitors that can be coupled in series with the first andsecond inductors 41 a and 41 b. For example, as the switching network 14can include N switching circuits SC[N:0] that each include a switch andone or more capacitive circuit elements. Each of the plurality ofcapacitors can be coupled in parallel with each other to increase theeffective capacitance of the LC tank. Control signals can toggleswitches in the switching circuits SC[N:0] to add and/or removeadditional capacitance from the effective capacitance of the LC tank,which can represent the combined capacitance of the tunable capacitanceelements that are part of the LC tank circuit. For instance, eachcapacitor of the capacitor switching network 14 circuit can beselectively included or excluded from the effective capacitance of theLC tank based on values of the capacitance control signals openingand/or closing switches, such as transistors. With additionalcapacitance, the oscillator frequency can decrease. Conversely, withreduced capacitance, the oscillator frequency can increase.

The oscillator 10 illustrated in FIG. 4 is a VCO. A tuning voltageV_(TUNE) can be applied to tuning elements 42 a and 42 b to apply changethe frequency of the LC tank. The tuning voltage V_(TUNE) can controlthe output frequency of the oscillator 10.

The passive impedance network 20 c can include a first capacitor 44 a, asecond capacitor 44 b, a first biasing inductor 46 a, and a secondbiasing inductor 46 b. The passive impedance network 20 c can apply abias voltage Vg to gates of transistors Mn and Mp so as to reduce thevoltage at the gates of transistors Mn and Mp. The first capacitor 44 acan have a first end coupled to the first node N1 and a second endcoupled to the gate of the transistor Mp. The first capacitor 44 a canhave a capacitance selected to block a DC voltage at the first node N1from being applied to the gate of the transistor Mp. The secondcapacitor 44 b can have a first end coupled to the second node N2 and asecond end coupled to the gate of the transistor Mn. The secondcapacitor 44 b can have a capacitance selected to block a DC voltage atthe second node N2 from being applied to the gate of the transistor Mn.The first biasing inductor 46 a can have a first end coupled to acircuit element driving the gate biasing voltage Vg and a second endcoupled to the gate of the transistor Mp. Similarly, the second biasinginductor 46 b can have a first end coupled to a circuit element drivingthe gate biasing voltage Vg and a second end coupled to the gate of thetransistor Mn. The first biasing inductor 46 a and the second biasinginductor 46 b can have inductances suitable to provide a low noise, highimpedance at a resonant frequency of the oscillator 10.

The sustaining amplifier of the oscillator 10 illustrated in FIG. 4includes transistors Mn and Mp. Transistors Mn and Mp can implement oneor more features of any of the sustaining amplifiers described herein.

The oscillator 10 can include a tail inductor 48. The tail inductor 48can have a first end coupled to the source of at least one transistor Mnand/or Mp of the sustaining amplifier and a second end coupled to adrain of a transistor Mb configured to provide a bias current to thesustaining amplifier. The inductance of the tail inductor 48 can beselected such that the tail inductor 48 is configured to resonate at adesired frequency and block a single-ended path to AC ground at thedesired frequency. For example, in some implementations, the tailinductor 48 can to configured to resonate the parasitic capacitance at acommon source node of the sustaining amplifier at twice the resonantfrequency ω of the oscillator 10. This can effectively increase theimpedance of the bias current source at the second harmonic. As aresult, the second harmonic of the thermal noise current may not have asingle-ended path to the AC ground. The tail inductor 48 can be includedin the bias source 18 of FIGS. 3A and/or 3B.

The bias source of the oscillator 10 illustrated in FIG. 4 also includesa bias transistor Mb and a bias capacitor Cb. The bias transistor Mb canhave a gate coupled to a bias voltage Vbias. The bias transistor Mb canbe configured to drive a current from a power rail (for example, Vdd) toa common node of the sustaining amplifier. As such, the bias transistorcan be configured as a current source. The bias capacitor Cb can have afirst end coupled to a source of the bias transistor Mb and a second endcoupled to the drain of the bias transistor. The bias capacitor Cb canfilter out noise from the bias source transistor.

Although three example passive impedance networks are provided in FIGS.3A, 3B, and 3C for illustrative purposes, it will be understood that anumber of other passive impedance can implement one or more aspects ofthe present disclosure to thereby reduce phase noise generated by asustaining amplifier. Likewise, although n-type or p-type transistorsare shown in FIGS. 3A, 3B, 3C, and 4 for illustrative purposes, it willbe understood that the illustrated oscillators can be modified to themirror image configuration by swapping n-type and p-type transistors,reversing currents, and swapping voltage rails. For example, in FIGS.3A-3C the illustrated sustaining amplifiers include NMOS devices Mn andMp and in FIG. 4 the illustrated sustaining amplifier includes PMOSdevices Mn and Mp. In some embodiments, an oscillator can include twosustaining amplifiers, one with n-type devices and one with p-typedevices. One or more of these two sustaining amplifier can be biased bya passive impedance network that includes any combination of featuresdescribed with reference to the passive impedance networks describedherein.

FIG. 5 shows graphs illustrating a relationship among transconductancesand current at the drains of differential transistors in a sustainingamplifier that indicate a reduction in zero crossing noise, according toan embodiment. The graphs of FIG. 5 show the relationship amongtime-domain waveforms of an oscillator with a passive impedance networksimilar to the passive impedance network 20 c of FIG. 4 and a similaroscillator without a passive impedance network. The graphs of FIG. 5illustrate sustaining amplifier transistor drain currents andtransconductances for transistors Mn and Mp of a sustaining amplifier(for example, as shown in FIGS. 3A, 3B, or 4). The top two graphs aresimulation results of a VCO that is functionally similar to theoscillator of FIG. 4, and the bottom two graphs are simulation resultsof a similar VCO without a passive impedance network.

With continued reference to FIG. 5, the drain current of transistor Mnis represented by the curve 51 and the drain current of transistor Mp isrepresented by the curve 52 for the oscillator with the passiveimpedance network. The transconductance of transistor Mn is representedby the curve 53 and the transconductance of transistor Mp is representedby the curve 54 for the oscillator with the passive impedance network.The drain current of transistor Mn is represented by the curve 55 andthe drain current of transistor Mp is represented by the curve 56 forthe oscillator without the passive impedance network. Thetransconductance of transistor Mn is represented by the curve 57 and thetransconductance of transistor Mp is represented by the curve 58 foroscillator without the passive impedance network.

In FIG. 5, a time period Tgds_(ENH) in which the sustaining amplifiertransistors of the oscillator with the passive impedance network stay onduring the zero crossing instant of the oscillator output issignificantly reduced compared to a time period Tgds_(CONV) in which thesustaining amplifier transistors of the oscillator without the passiveimpedance network stay on during the zero crossing instant. As shown inFIG. 5 and summarized in Table 1 below, the time period in whichtransistors in sustaining amplifiers are on can be reduced by a factorof about 4.5 times or more, in some implementations. This can alsocorrespond to a reduction of thermal noise current injected into the LCtank at the zero crossing instants. More specifically, according to thesimulation results shown in FIG. 5, the amount of RF current Izc_(ENH)injected in the oscillator with the passive impedance network should beabout 5 times less than the thermal noise current Izc_(CONV) injected inthe oscillator without the passive impedance network. The reduction inthe time in which sustaining amplifier transistors stay on during thezero crossing instant of the oscillator output can translate into asignificant reduction in the thermal noise injected into the LC tank atthe sensitive zero-crossing instants, which in turn can result in areduction in the oscillator phase noise.

The simulation results shown in FIG. 5 also indicate that the period oftime Ti_(ENH) in which the sustaining amplifier transistors of theoscillator with the passive impedance network operate in the linearregion should be about 1.4 times less than the time Ti_(CONV) in whichthe sustaining amplifier transistors of the oscillator without thepassive impedance network operate in the linear region. This canrepresent a reduction in the amount of time in which the sustainingamplifier transistors load the oscillator LC tank and hence a reductionin the noise contributed by this resistive loading to the thermal noisein the 1/f² region of the oscillator phase noise.

Table 1 below summarizes measurements of Ti and Tgds derived from thegraphs shown in FIG. 5.

TABLE 1 Ti_(ENH) 0.015 ns Ti_(ConV) 0.070 ns Tgds_(ENH) 0.081 nsTgds_(ConV) 0.115 ns

FIGS. 6A and 6B are graphs illustrating relationships among phase noiseand frequency that show improved noise performance for oscillators withpassive impedance networks configured to bias sustaining amplifiers.These graphs show comparative simulations illustrating the improvementin performance achieved by oscillators with passive impedance networkscompared to oscillators without passive impedance networks in terms ofphase noise at roughly the same bias current conditions. A firstoscillator with the passive impedance network in these simulations is aVCO that is functionally similar to the oscillator of FIG. 4, and asecond oscillator without the passive impedance network is similar tothe oscillator of FIG. 4 with the exception of the passive impedancenetwork.

FIG. 6A shows a phase noise comparison between the first oscillator andthe second oscillator for a lowest frequency band in both high and lowbias conditions. In contrast, FIG. 6B shows a phase noise comparisonbetween first oscillator and the second oscillator for a highestfrequency band for both high and low bias conditions. The simulatedoscillators each have 64 frequency bands that can be selected by sixseparate switching circuits in the switching network 14 of FIG. 1. Morespecifically, in FIG. 6A, the phase noise of the lowest frequency bandunder high bias conditions in the first oscillator is represented bycurve 61, the phase noise of the lowest frequency band under low biasconditions in the second oscillator is represented by curve 62, thephase noise of the lowest frequency band under high bias conditions inthe second oscillator is represented by curve 63, the phase noise of thelowest frequency band under low bias conditions in the second oscillatoris represented by curve 64. In FIG. 6B, the phase noise of the highestfrequency band under high bias conditions in the first oscillator isrepresented by curve 65, the phase noise of the highest frequency bandunder low bias conditions in the first oscillator is represented bycurve 66, the phase noise of the highest frequency band under high biasconditions in the second oscillator is represented by curve 67, thephase noise of the highest frequency band under low bias conditions inthe second oscillator is represented by curve 68.

Table 2 shows that for the same bias current at the lowest bias currentsetting in both the first oscillator and the second oscillator, theoscillator RMS fundamental voltage is higher for the first oscillatorthat includes a passive impedance network. This can be a result of thereduction in time in which the enhanced oscillator sustaining amplifiertransistors operate in the linear region. The phase noise improvementachieved by using the first oscillator shown by FIGS. 6A and 6B is about4.6 dBc/Hz for the lowest band setting and 5.6 dBc/Hz for the highestband setting at 1 MHz frequency offset from the RF carrier.

TABLE 2 Oscillator without Oscillator with Oscillator without Oscillatorwith Passive Impedance Passive Impedance Passive Impedance PassiveImpedance Network (highest Network (highest Network (lowest Network(lowest frequency band, frequency band, frequency band, frequency band,low bias) low bias) low bias) low bias) Oscillator Fundamental  3.45 GHz 3.41 GHz  3.03 GHz  3.05 GHz Frequency Oscillator Core Current 29.99 mA29.46 mA 35.88 mA 35.46 mA Consumption Oscillator Core RMS 3.58 V  4.63V  3.32 V  4.17 V  Fundamental Voltage of Carrier Relative Difference inRMS   0 dB  2.25 dB   0 dB  2.86 dB Amplitude Total Phase Noise (dBc/Hz) 0.4 MHz −129.86 −133.97 −130.63 −136.11  1 MHz −138.00 −142.60 −138.63−144.16  10 MHz −155.30 −157.79 −155.34 −157.88 100 MHz −158.45 −159.29−157.99 −158.96

Table 3 shows a comparison between the first oscillator and the secondoscillator at a high current setting. At the high current setting inwhich the oscillator is supply voltage limited, hard gate oxidebreakdown and/or hot carrier effects can limit operation. Simulationindicates that the first oscillator with a passive impedance network canachieve 3.5 dBc/Hz better phase noise at 1 MHz frequency offset from theRF carrier for the lowest band setting and 5.6 dBc/Hz better phase noisefor the highest band compared to the second oscillator without thepassive impedance network.

TABLE 3 Oscillator without Oscillator with Oscillator without Oscillatorwith Passive Impedance Passive Impedance Passive Impedance PassiveImpedance Network (highest Network (highest Network (lowest Network(lowest frequency band, frequency band, frequency band, frequency band,high bias) high bias) high bias) high bias) Oscillator Fundamental  3.44GHz  3.38 GHz  3.03 GHz  3.04 GHz Frequency Oscillator Core Current44.22 mA 47.73 mA 47.53 mA 51.02 mA Consumption Oscillator Core RMS 4.78V  5.15 V  4.32 V  4.72 V  Fundamental Voltage of Carrier RelativeDifference in RMS   0 dB  0.65 dB   0 dB  0.77 dB Amplitude Total PhaseNoise (dBc/Hz)  0.4 MHz −131.92 −134.93 −132.57 −136.70  1 MHz −140.00−143.36 −140.52 −144.65  10 MHz −156.76 −158.16 −156.71 −158.39 100 MHz−159.39 −159.57 −158.96 −159.50

Reducing Noise Generated by Switching Network

Phase noise of a second source can be generated by a switching networkconfigured to tune a resonant circuit, such as an LC tank, to a desiredresonant frequency. A switch configured to switch in and/or switch outcircuit elements, such as capacitors, to vary the resonant frequency canbe biased such that nodes of the switch are not floating when the switchis off. When the switch is off, a high impedance asserted on nodes ofthe switch can reduce contributions of the switching circuit to thephase noise of the oscillator.

However, generating a high impedance when the switch is off can beexpensive to implement via large resistors, for example, resistorshaving a resistance from about 100 k Ohms to about 150 k Ohms. Inswitching circuits (for example, the circuit illustrated in FIG. 7), theparasitic capacitance of such large resistors can affect the ratio of acapacitance when the switch is on or off. This can, for example, reducethe change in resonant frequency of an LC tank when a capacitor isswitched in or switched out the LC tank.

FIG. 7 is a block diagram of a switching network 14. The switchingnetwork 14 can selectively couple circuit elements across a first nodeand a second node of a resonant circuit, such as nodes N1 and N2 of theresonant circuit 12 of FIG. 1, to thereby increase and/or decrease theresonant frequency of a resonant circuit. For example, in someimplementations, the switching network can selectively couple capacitorsacross an LC tank to increase and/or decrease the effective capacitanceof the LC tank. Adjusting the effective capacitance of the LC tank canadjust the resonant frequency ω of the LC tank, for example, based onEquation 1. A switch, such as a field effect transistor, can beconfigured to switch-in or switch-out capacitors across the LC tank.

The switching network 14 can include a switching network driver 72 andan array 74 of switching circuits 76 a to 76 n. The array 74 can beconfigured to selectively couple circuit elements to a first node N1 anda second node N2 of the resonant circuit to adjust the resonantfrequency ω of the resonant circuit 12 to a selected frequency band. Thefirst node N1 and the second node N2 can be a non-inverted node and aninverted node, respectively. Control signals Control [2N-1:0] can begenerated by the switching network driver 72 to turn switches ofswitching circuits 76 a to 76 n on and/or off. At least some of controlsignals Control [2N-1:0] can select a frequency band by turning onselected switches within the switching circuits 76 a to 76 n. Thesignals that select the frequency band can be referred to as bandcontrol signals.

The switching network driver 72 can include a level shifter and an arrayof buffer drivers. The level shifter can generate a control bias voltageto control the switches and an intermediate node bias voltage to biasother nodes of the switches. For instance, the control bias voltage canbe coupled to a gate of a field effect transistor in a switchingcircuit, and the intermediate node bias voltage can be coupled to asource and a drain of the field effect transistor in the switchingcircuit. The level shifter can adjust the voltage level of a supplyvoltage (for example, a battery voltage) to the control bias voltage andthe intermediate node bias voltage. In some implementations, an off-chipcapacitor can filter out noise on one or more of the bias voltages. Thebias voltages can be used as logical high voltages for devices in thearray of buffer drivers. The array of buffer drivers can drive thecontrol bias voltage and the intermediate node bias voltage to switchingcircuits 76 a to 76 n. Using the level-shifted high voltage valuesgenerated by the level shifter can avoid breakdown of the junctions ofthe switches in the switching circuits 76 a to 76 n.

Each switching circuit 76 a to 76 n can receive at least one bandcontrol signal to turn a switch on or off. When the switch is on, atleast one circuit element, such as a capacitor, can be coupled to thefirst node N1 (for example, a non-inverted node) and a second node N2(for example, an inverted node) of the oscillator. For example, in an LCtank implementation, when the switch is on, a first end of a capacitorcan be coupled to node N1 and a second end of the capacitor can becoupled via the switch to the second node N2. This coupling can increasethe capacitance of the LC tank by adding capacitance in parallel,thereby adjusting the resonant frequency ω of the LC tank. Conversely,when the switch is off, at least one circuit element, such as acapacitor, can be uncoupled from at least one of the first node N1 andthe second node N2 of the oscillator. For example, in an LC tankimplementation, a capacitor can be decoupled from node N1 and node N2when the switch is off.

When a switch of one of the switching circuits 76 a to 76 n is on, theimpedance of the switch may not introduce significant phase noise to theresonant circuit. Yet, when the switch is off, the second bias voltagecan be applied across the switch such that the nodes across the switchdo not have undefined voltage. For instance, a source and a drain of afield effect transistor can be biased with the second bias voltage suchthat the source and the drain do not float.

When the switch is off, a high impedance can be desired to switch outthe capacitor from the LC tank. However, when the switch is off, aparasitic capacitance of the switch and/or additional circuitry in theswitching circuits 76 a to 76 n can add capacitance to the LC tank.Additional capacitance added to the effective capacitance of the LC tankcan reduce the effectiveness of switching in or switching out acapacitor. Larger switch sizes can exacerbate problems related to theparasitics of the switch when the switch is off. Accordingly, a highimpedance and low parasitic capacitance when the switch is off canswitch out a capacitor without having the parasitic capacitance of theswitch significantly effect the effective capacitance of the LC tank.

Integration of passive circuit elements (for example, large resistors)sufficient to generate a high impedance coupled to the source and/ordrain of a switch in a low cost process can be prohibitive. Forinstance, resistors having an impedance of about 100 kOhm to 150 kOhmcan be quite costly. Sensitive circuit areas, as around the RF bandswitches (for example, switches 84 that will be described with referenceto FIGS. 8A, 8B, and/or 9) where the parasitic capacitances introducedas a result of using such a large resistor can be detrimental to theon/off capacitance ratio of the band switch can be problematic.

FIG. 8A is a schematic diagram of a resonant circuit 12A including oneswitching circuit, according to an embodiment. The switching circuit isone example of a switching circuit 76 a to 76 n of FIG. 7. Although oneswitching circuit is shown for illustrative purposes, any suitablenumber of switching circuits can be included in series and/or parallelwith the switching circuit illustrated in FIG. 8A. The switching circuitcan include a first active circuit 82 a, a second active circuit 82 b, aswitch 84, a first circuit element 86 a, and a second circuit element 86b. The switching circuit can be coupled to the first node N1 and thesecond node N2 of a resonant component 88 of the resonant circuit 12A.

The first active circuit 82 a can be coupled to a node S1 that isintermediate a first end of the switch 84 and the first node N1. Thenode S1 can also be intermediate a first end of the switch 84 and afirst end of the first circuit element 86 a, for example, as shown inFIG. 8A. A second end of the first circuit element 86 a can be coupledto the first node N1 of the resonant component 88. Similarly, the secondactive circuit 82 b can be coupled to a node S2 intermediate a first endof the switch 84 and the second node N2. In some implementations, thesecond active circuit 82 b can be coupled to a node S2 intermediate asecond end of the switch 84 and a first end of the second circuitelement 86 b. A second end of the second circuit element 86 b can becoupled to the second node N2 of the resonant component 88.

The first active circuit 82 a and the second active circuit 82 b includeactive circuit elements configured to deliver bias voltage. Non-limitingexamples of active circuit elements include transistors, diodes, and thelike. The first active circuit 82 a and/or the second active circuit canreceive two bits of the control signals Control [2N-1:0], for example,from the switching network driver 72. One of these two control bits canbe used to select a frequency band of operation and the other controlbit can be used to apply a bias on an intermediate node, such as node S1and/or node S2.

The first active circuit 82 a can generate an impedance on the node S1.When the switch 84 is off, the first active circuit 82 a can generate ahigh impedance on the node S1. The first active circuit 82 a cangenerate the high impedance based on a bias signal with a differentlogical high value than a band control signal configured to control theswitch. For example, the high impedance can be generated using theintermediate node bias signal described in reference to FIG. 7 and theswitch can open and/or close based on the control bias signal describedin reference to FIG. 7. The high impedance can be sufficient toeffectively create an open circuit between nodes S1 and S2. Forinstance, in some implementations, the active circuit 82 a can generatean impedance of at least about 100 k Ohms, 150 k Ohms, 1 M Ohm, 1 G Ohm,or more. When the switch 84 is on, the active circuit 82 a can stopgenerating the high impedance on node S1. Alternatively or additionally,the active circuit 82 a can pull down node S1 while the switch is on.For instance, the active circuit 82 a can pull down node S1 based on aband control signal configured to open and/or close the switch 84.

The second active circuit 82 b can include and/or implement anycombination of features of the active circuit 82 a. Where the firstactive circuit 82 a is coupled to node S1, the second active circuit 82b is correspondingly coupled to the node S2.

According to some implementations, any of the active circuits 82 aand/or 82 b can be replaced by a passive circuit that includes aninductor configured to apply a bias voltage in place of the activecircuit 82 a and/or 82 b.

The switch 84 can be any suitable voltage controlled switch. Forexample, the switch 84 can be a field effect transistor. The switch canopen and/or close in response to a control signal, such as a bandcontrol signal. The switch 84 can couple the second end of the firstcircuit element 86 a to the second node N2 of the resonant component 88when on. The switch 84 can also couple the second end of the secondcircuit element to the first node N1 of the resonant component 88 whenon. By selectively coupling the first circuit element 86 a and/or thesecond circuit element 86 b across the resonant circuit 88, the switch84 can adjust the resonant frequency of the resonant circuit 88.

The resonant component 88 can include any circuit configured tooscillate at a resonant frequency. When the switch 84 is on, theillustrated switching circuit can be considered part of the resonantcomponent 88. When the switch 84 is off, the illustrated switchingcircuit should not be considered part of the resonant component 88. Insome implementations, the resonant component 88 can include an LC tankwith a capacitor Ctank in parallel with an inductor Ltank. The capacitorCtank can represent one or more capacitors in series and/or parallel.Likewise, the inductor Ltank can represent one or more inductors inseries and/or parallel.

FIG. 8B is a schematic diagram of a resonant circuit 12B including oneswitching circuit, according to another embodiment. Like the resonantcircuit 12A, any suitable number of switching circuits can be includedin parallel and/or in series in the resonant circuit 12B. The resonantcircuit 12B of FIG. 8B can be substantially the same as the resonantcircuit 12A of FIG. 8A, except that the resonant circuit 12B includes asingle circuit element 86 instead of the first circuit element 86 a andthe second circuit element 86 b.

FIG. 9 is a schematic diagram of a switching circuit 76 according to anembodiment. The switching circuit 76 illustrated in FIG. 9 is an exampleswitching circuit 76 a that can be implemented in one or more of theswitching circuits 76 a to 76 n of the array 74 of switching circuitsillustrated in FIG. 7. One switching circuit 76 can be included for eachfrequency band of an oscillator. In some implementations, 2, 4, 8, 16,32, 64, 128 or more switching circuits 76 can be included in an array.The switching circuit 76 can include a first active circuit 82 a, asecond active circuit 82 b, a switch 84, a first circuit element 86 a, asecond circuit element 86 b, or any combination thereof.

The first active circuit 82 a of FIG. 9 is one example of the firstactive circuit 82 a of FIGS. 8A and/or 8B. Likewise, the second activecircuit 82 b of FIG. 9 is one example of the second active circuit 82 bof FIGS. 8A and/or 8B. Any combination of features of the activecircuits 82 a and/or 82 b described herein can be implemented in anyother active circuits described herein.

Each of the active circuits illustrated in FIG. 9 include a pull-uptransistor 90, 92 and a pull-down transistor 94, 96. The pull-uptransistors 90, 92 can be field effect transistors, such as PMOStransistors. In some implementations, the pull-up transistor 90 and/or92 can be diode connected, as shown in FIG. 9. The pull-up transistor 92can receive a bias voltage Control[0], such as the intermediate nodebias voltage described with reference to FIG. 7, and apply the biasvoltage Control[0] to a node S1 between the drain of the switch 84 and afirst end of the first circuit element 86 a. Similarly, the pull-uptransistor 92 can receive the bias voltage Control[0] and apply the biasvoltage Control[0] to a node S2 between the source of the switch 84 anda first end of the second circuit element 86 b.

When the switch 84 is off, the nodes S1 and/or S2 can be at a potentialof about Vhigh-Vth, in which Vhigh can represent a logical high valueand Vth can represent the threshold voltage of the pull-up transistors90 and/or 92. The diode connected transistors 90 and/or 92 can operatein a saturation mode. In the saturation mode, the diode connectedtransistors 90 and 92 can provide a high DC impedance on nodes S1 andS2, respectively. In addition, the diode connected transistors 90 and 92can each generate an impedance on the order of 100s of Mega Ohms at highfrequencies (for example, RF frequencies). The diode connectedtransistors 90 and 92 do not add significant parasitic capacitance onnodes S1 and/or S2 or other nodes of the switching circuit 76. The diodeconnected transistors 90 and 92 can be configured to turn off when theswitch 84 is on. For example, the bias voltage Control[1] can be thelogical compliment of the voltage applied to the gate of the switch 84.

When the switch 84 is on, pull down transistors 94 and 96 can pull downnodes S1 and S2, respectively. In some implementations, the pull downtransistors 94 and 96 can be controlled by the same signal applied tothe gate of the switch 84. The pull down transistors 94 and 96 can besmall in size relative to the switch 84. Consequently, the pull downtransistors 94 and 96 may not add significant parasitic capacitance onnodes S1 and/or S2 or other nodes of the switching circuit 76.

The impedance generated by the active circuits 82 a and 82 b on nodes S1and S2, respectively in combination with the parasitic capacitances atnodes S1 and S2 can form a noise filter with a relatively low cornerfrequency. Accordingly, the active circuits 82 a and 82 b can reducenoise in the switching circuit 76.

In the implementation illustrated in FIG. 9, the first circuit element86 a and the second circuit element 86 b are differential capacitors.These capacitors can adjust the resonant frequency of a resonantcircuit, such as an LC tank, for example, as described above.

Simulation results indicate that noise voltage spectral density of acapacitor switching circuit with resistors biasing source and drain of aswitch 84 in the off-state can increase by 10 dB for every decadeincrease in impedance across source and drain of a field effect switch84 when the switch 84 is off. A first-order low pass filter can beformed by the impedance across the source and drain of the field effectswitch 84 and the parasitic capacitances formed by the gate and sourceof the switch 84, the gate and drain of the switch 84, and passivecircuits, such as resistor networks, coupled to nodes S1 and/or S2.Thus, for every decade increase in impedance across source and drain ofa field effect switch 84, there can also be a decade decrease in thefirst-order low-pass filtering of noise in the capacitor switchingnetwork in capacitor switching circuits with resistors biasing thesource and drain of the switch 84. This can attenuate the noisecontribution of the switch 84 in the off-state, particularly at highfrequencies in the GHz range. Other simulation results indicate thatimpedance of circuits configured to bias nodes S1 and/or S2 candetermine the low-pass noise filter corner frequency.

FIGS. 10A and 10B are graphs illustrating relationships among noisevoltage spectral density in switching circuits showing a reduction innoise generated by a switching circuit, according to an embodiment. FIG.10A shows a graph of the noise voltage spectral density of a capacitorswitching network in the off-state (i.e., when the switch 84 is off)that compares active circuits and passive resistor-based circuitscoupled to intermediate nodes S1 and S2. Curve 102 represents the noisevoltage spectral density of a switching circuit with active circuits,which are functionally similar to the active circuits 82 a and 82 b, ofFIG. 9 coupled to nodes S1 and S2, respectively. In contrast, curve 104represents the noise voltage spectral density of a switching circuitpassive resistor-based circuits configured to bias nodes S1 and S2, withthe resistive circuits having an impedance of about 30 k Ohm from sourceto drain of a field effect switch 84. FIG. 10A shows that AC noisevoltage spectral density is reduced by about 9 dB at 3 GHz in the curve102 compared to the curve 104.

FIG. 10B shows graph comparing phase noise of a resistor pull-up/pulldown network and a PMOS switch based network in a switching circuit.Curve 106 represents phase noise of a first VCO that includes a passiveresistor-based circuit configured to bias intermediate nodes S1 and S2of a band-switch circuit. Curve 108 represents phase noise of a secondVCO that includes a PMOS-based circuit configured to bias intermediatenodes S1 and S2 of a band-switch circuit. Table 4 summarizes data fromthe curves 106 and 108 in the graph of FIG. 10B. The data indicate thatperformance parameters, such as the fundamental frequency, currentconsumption and amplitude are roughly the same for the first VCO and thesecond VCO. The data indicate that phase noise of the first VCO and thesecond VCO is roughly the same when all the band-switches are switchedin across an LC tank (i.e., at the lowest fundamental frequency) asexpected, since the phase noise appears to be dominated by the onresistance of the band-switch. However, the data in Table 4 indicateimprovements in phase noise of the second VCO compared to the first VCOof about a 1.5 dB at frequency offsets of about 0.4 MHz to 2 MHz andabout a 1 dB improvement in phase noise at low frequency offsets ofabout 100 kHz. The 1.5 dB improvement in phase noise can result fromeliminating the contribution of the pull-up resistors in theband-switches to the total phase noise, when the band-switch is off. Thecontributions to phase noise listed in Table 4 indicate that thecontribution of the pull-up resistors in the band-switches in the firstVCO to the total phase noise is about 29%.

TABLE 4 R-Based R-Based Active Active (Highest (Lowest (Highest (LowestBand) Band) Band) Band) VCO  3.80 GHz  3.22 GHz  3.78 GHz  3.22 GHzFundamental Frequency VCO Core 19.59 mA 30.40 mA 19.61 mA 30.55 mACurrent Consumption VCO 3.33 V  3.81 V  3.31 V  3.78 V  Core RMS VoltageTotal phase noise (dBc/Hz) at  0.1 MHz −115.74 −119.82 −116.91 −119.44 1 MHz −136.57 −140.45 −138.15 −140.41  10 MHz −155.95 −158.10 −156.41−157.94 100 MHz −160.93 −159.75 −160.92 −161.38 Phase 28.89 NegligibleNoise Power from Pull-Up Devices (%) Phase Noise 47.08 74.04 Power fromSustaining amplifier (%) Phase Noise 12.46 18.31 Power from inductor (%)

FIGS. 11A and 11B are graphs illustrating voltage swings in switchingcircuits showing that voltage swings stay within a desired range ofbreakdown voltages according to an embodiment. The simulation switchingcircuits are functionally similar to the active circuits 82 a, 82 b ofFIG. 9. In FIG. 11A, curve 112 in a time domain waveform of voltage atthe source of the switch 84 of FIG. 9 and curve 113 is a time domainwaveform of voltage at the drain of the switch 84 of FIG. 9. Thegate-to-source voltage is represented by curve 114 and the gate-to-drainvoltage is represented by the curve 115. FIG. 11A corresponds to highbias/maximum bias conditions. Similar results were observed across therange of frequency bands for the VCO.

FIG. 11B shows simulation results from the same simulation correspondingto FIG. 11A, except that FIG. 11B includes data from low/minimum biasconditions. In FIG. 11B, curve 116 in a time domain waveform of voltageat the source of the switch 84 of FIG. 9 and curve 117 is a time domainwaveform of voltage at the drain of the switch 84 of FIG. 9. Thegate-to-source voltage is represented by curve 118 and the gate-to-drainvoltage is represented by the curve 119. FIG. 11A corresponds to highbias/maximum bias conditions. Similar results were observed across therange of frequency bands for the VCO.

The curves shown in FIGS. 11A and 11B indicate that the voltage swingsshould not exceed the rated breakdown voltages at any of the junctionsby more than 10%. This is within acceptable bounds.

Other data indicate that a capacitor switching with active circuits 82 aand 82 b (for example, as shown in FIG. 9) and the passive impedancenetwork 20 c (for example, as shown in FIG. 4) can lead to about 5 dBimprovement in phase noise of a VCO. The active circuits 82 a and 82 bcan improve phase noise by about 2.5 dB according to other data. Thepassive impedance network 20 c can improve phase noise in the sustainingamplifier by about 3 dB according to the other data.

With one or more of the improvements in phase noise described herein,VCOs can meet challenging noise requirements. For instance, simulationresults indicate that VCOs manufactured with a SiCMOS process with oneor more features described herein can meet the Multi-Carrier GSM TX 1800noise specification.

Table 5 shows some of the phase noise data for an oscillator with apassive impedance network 20 c of FIG. 4 and switching circuits 76 ofFIG. 9. The data in Table 5 indicate that the phase noise of an enhancedVCO with the passive impedance network 20 c and switching circuits 76can improve phase noise by about 2.6 dB compared to some conventionalVCOs. The enhanced VCO can have about −137.1 dBc/Hz phase noise at 1 MHzoffset from a 3.3 GHz RF carrier.

TABLE 5 Core Enhanced VCO Conventional VCO Phase Noise at 1 MHz HighestBand: −134.7 Highest Band: −132.2 (dBc/Hz) (VCO Bias = 32) (VCO Bias =3) Vtune = 1.5 V, Lowest Band: −137.1 Lowest Band: −135 Vdd = 2.7 V (VCOBias = 32) (VCO Bias = 3) Tuning Range (MHz) 3682-3284 3613-3172 (398,12%) (441, 13.9%) Correction Factor with Highest Band: 0 +0.16 respectto Center Lowest Band: 0 +0.3 Frequency (dB) Effective Phase NoiseHighest Band: 0 +2.66 degradation with Lowest Band: 0 +2.40 respect toenhanced VCO (dB)

Reducing Noise Generated by Sustaining Amplifier and Switching Network

In some embodiments, oscillators can be configured to reduce the noisegenerated by a sustaining amplifier and noise generated by the switchingnetwork. Any combination of features described with reference to FIGS.3A, 3B, or 4 can be implemented in concert with any combination offeatures described with reference to FIGS. 7, 8A, 8B, or 9.

For example, an oscillator can include a resonant circuit, a sustainingamplifier, a passive impedance network, and a switching network. Theresonant circuit can have a first end and a second end. In someimplementations, the first end and the second end are a non-invertednode and an inverted node, respectively. The sustaining amplifier caninclude a first switch configured to drive the first end of the resonantcircuit in response to an input at a first control terminal of the firstswitch and a second switch configured to drive the second end of theresonant circuit in response to an input at a second control terminal ofthe second switch. The passive impedance network can include one or moreexplicit passive impedance elements. The passive impedance network canbe configured to pass a bias to the first control terminal of the firstswitch and the second control terminal of the second switch. Theswitching network can include one or more switching circuits that areconfigured to tune a resonant frequency of the resonant circuit. Each ofone or more switching circuits can include a circuit element, a switch,and an active circuit. The circuit element can have at least a first endand a second end. In some implementations, the circuit element can be acapacitor. The switch can be configured to selectively couple the secondend of the circuit element to the second end of the resonant circuit.The active circuit can be configured to assert a high impedance on anintermediate node between the switch and the first node when the switchis off. For instance, the high impedance can be asserted in response tothe switch turning off. In some implementations, the intermediate nodeis between the switch and the second end on the circuit element. Thepassive impedance network can effectively be in parallel with theresonant circuit. Accordingly, the passive impedance network and thecircuit element of the switching circuit can both contribute to settingthe resonant frequency of the resonant circuit. For instance, theinductance and/or capacitance of the passive impedance network combinedwith a capacitance of a capacitor of one or more switching circuitcoupled in parallel with the resonant circuit can contribute to settingthe resonant frequency.

Secondary LC Tuning Network

The passive impedance networks in the oscillators described above can beLC networks. For instance, the passive impedance network 20 c isillustrated as an LC network in FIG. 4 and one or more the of thepassive impedance networks 20 a of FIG. 3A, 20 b of FIGS. 3B, and 20 cof FIG. 3C can be LC networks. In certain embodiments, a tuning networkcan be added to any of the LC networks described above. The secondary LCtuning network can be implemented in combination with one or more of thefeatures described herein.

The tuning network can provide differential tuning between the controlterminals of opposing switches in a differential pair of switches, suchas the switches Mn and Mp described above, of a sustaining amplifier ofan oscillator, according to some embodiments. The tuning network canadjust the impedance between the control terminals of opposing switchesof the differential pair of switches in the sustaining amplifier. Forinstance, the tuning network can adjust a capacitance between thecontrol terminals of the differential pair of switches. The differentialpair of switches can be a differential pair of field effect transistors.

In some other embodiments, the tuning network can adjust an impedance inan LC tuning network electrically coupled to a control terminal of aswitch, such as a gate of a field effect transistor, of a single endedsustaining amplifier. For example, the tuning network can adjust acapacitance that is electrically coupled to the control terminal of thesingle ended sustaining amplifier.

According to some embodiments, the tuning network can extend a tuningrange of an oscillator. Alternatively or additionally, the secondary LCnetwork including the tuning network can reduce or eliminate a need fora tail inductor of the oscillator, such as the tail inductor 48 of FIG.4, according to certain embodiments. The secondary LC network includingthe tuning network can reduce an amount of time that a field effecttransistor of the differential pair operates in the Ohmic region in someembodiments.

FIG. 12 is a schematic diagram of an oscillator 100 having a secondaryLC tuning network 20 d, according to an embodiment. As illustrated, thesecondary LC tuning network 20 d is configured to bias control terminalsof a sustaining amplifier that includes a differential pair oftransistors Mn and Mp. Transistors of the differential pair oftransistors are shown as p-type transistors in FIG. 12. The field-effecttransistors (FETs) or “transistors” described herein can correspond totransistors known as metal-oxide-semiconductor field-effect transistors(MOSFETs). While the terms “metal” and “oxide” are present in the nameof the device, it will be understood that these transistors can havegates made out of materials other than metals, such as polycrystallinesilicon, and can have dielectric “oxide” regions made from dielectricsother than silicon oxide, such as from silicon nitride or high-kdielectrics.

The oscillator 100 includes a resonant circuit having a first node N1and a second node N2. The signals at the first node N1 and the secondnode N2 can be sinusoidal signals that are 180 degrees out of phase witheach other, in some implementations. For instance, the first node N1 andthe second node N2 can have voltages that have opposite signs andapproximately the same magnitude at any given time. In someimplementations, the first node N1 and the second node N2 can bereferred to as a non-inverted node and an inverted node, respectively,and the signals at these nodes can have values that are inverted fromeach other. The resonant circuit can operate at a resonant frequency.The resonant circuit of the oscillator 100 can be referred to as aprimary LC circuit when the resonant circuit includes an LC circuit. Theoscillator 100 illustrated in FIG. 12 includes a primary LC circuit thatincludes first and second inductors 41 a and 41 b, a capacitor switchingnetwork 14, and a capacitance 49. The oscillator 100 can oscillate, forexample, at a frequency selected from the range from about 3.4 GHz toabout 3.9 GHz. Simulation results indicate that the oscillator 100 canachieve a phase noise of about −140 dBc/Hz at a frequency offset ofabout 800 kHz relative to a 3.6 GHz carrier. Such an oscillator canconsume about 300 mW of power in some applications.

The inductive circuit element(s) of the primary LC circuit can includethe first and second inductors 41 a and 41 b. The first inductor 41 acan include a first end coupled to a power rail (for example, ground)and a second end coupled to the first node N1. The second inductor 41 bcan include a first end coupled to a power rail (for example, ground)and a second end coupled to the second node N2. The effective inductanceof the primary LC circuit can be based on inductance values of the firstinductor 41 a and the second inductor 41 b. In another implementation, asingle differential inductor with a central tap electrically connectedto a power rail, such as a ground potential or a power rail Vdd, can beused in place of the first inductor 41 a and the second inductor 41 b.

The effective capacitance of the primary LC circuit can be based oncapacitance values of the capacitor switching network 14. The capacitorswitching network 14 of the oscillator 100 can include any combinationof features of the capacitance switching networks discussed herein. Forinstance, the capacitance switching network 14 can include one or morefeatures of the capacitor switching network 14 of FIG. 4.

The oscillator 100 illustrated in FIG. 12 can be a VCO. A tuning voltageV_(TUNE) can be applied to tuning elements 42 a and 42 b to change thefrequency of the primary LC circuit. The tuning elements 42 a and 42 bcan be varactors as shown in FIG. 12. The tuning voltage V_(TUNE) cancontrol the output frequency of the oscillator 100.

The secondary LC tuning network 20 d can include a first capacitor 44 a,a second capacitor 44 b, a first biasing inductor 46 a, a second biasinginductor 46 b, and a tuning network 47. In addition to tuning as will bedescribed later, the secondary LC tuning network 20 d can apply a biasvoltage Vg that is provided as an input to the oscillator 100 to thegates of the transistors Mn and Mp so as to bias the voltage at thegates of transistors Mn and Mp. The bias V_(G) can be a voltage at apower rail voltage (for example, at a ground potential or at Vdd) insome implementations. The bias V_(G) can be at another suitable voltagelevel in some other implementations.

The first capacitor 44 a can be coupled in series between the first nodeN1 and the gate of the transistor Mp. The first capacitor 44 a can havea first end coupled to the first node N1 and a second end coupled to thegate of the transistor Mp. The first capacitor 44 a can have acapacitance selected to block a DC voltage at the first node N1 frombeing applied to the gate of the transistor Mp. The second capacitor 44b can be coupled in series between the second node N2 and the gate ofthe transistor Mn. The second capacitor 44 b can have a first endcoupled to the second node N2 and a second end coupled to the gate ofthe transistor Mn. The second capacitor 44 b can have a capacitanceselected to block a DC voltage at the second node N2 from being appliedto the gate of the transistor Mn.

The first biasing inductor 46 a can have a first end coupled to the gatebiasing voltage V_(G) and a second end coupled to the gate of thetransistor Mp. Similarly, the second biasing inductor 46 b can have afirst end coupled to the gate biasing voltage V_(G) and a second endcoupled to the gate of the transistor Mn. The first biasing inductor 46a and the second biasing inductor 46 b can have inductances suitable toprovide a low noise and/or high impedance at a resonant frequency of theoscillator 100. In another implementation, a single differentialinductor with a central tap electrically connected to a power rail, suchas a ground potential or Vdd, can be used in place of the first biasinginductor 46 a and the second biasing inductor 46 b.

The tuning network 47 can include a capacitor switching network. Forexample, the tuning network 47 can include any combination of featuresof the capacitor switching network 14 of FIG. 4 and/or the capacitorswitching network 14 of FIG. 7. In one embodiment, the tuning network 47can include one or more features of the switching circuits 76 of FIG. 7and/or FIG. 9. Control signals can toggle switches in the tuning network47 to add and/or remove capacitance from the effective capacitance ofthe secondary LC circuit that includes the tuning network 47 and thefirst and second biasing inductors 46 a and 46 b. With additionalcapacitance, the frequency at which the secondary LC circuit resonatescan decrease. Conversely, with reduced capacitance, the frequency atwhich the secondary LC circuit resonates can increase.

As illustrated in FIG. 12, the tuning network 47 is electrically coupledbetween the gates of the differential pair of transistors Mn and Mp ofthe sustaining amplifier. In this embodiment, the tuning network canincrease and/or decrease the capacitance between the gates of thedifferential pair of transistors Mn and Mp.

In the secondary LC tuning network 20 d, the first and second biasinginductors 46 a and 46 b can resonate the capacitance of the tuningnetwork 47 that is electrically coupled to the first and secondinductors 46 a and 46 b. By adjusting the capacitance that iselectrically coupled to the first and second inductors 46 a and 46 b,the tuning network 47 can increase the tunability of the primary LCcircuit. According to certain embodiments, the first and secondinductors 46 a and 46 b can resonate the capacitance of the sustainingamplifier coupled to the first and second inductors 46 a and 46 b so asto reduce phase noise of the oscillator and/or to reduce a conductionangle of the sustaining amplifier.

According to some embodiments, one or more tuning elements, such asvaractors, can be electrically coupled to the gate of the transistor Mnand/or the transistor Mp. These tuning elements can receive a secondarytuning voltage to tune the oscillation of the secondary LC network. Thesecondary tuning voltage can be different from the tuning voltageV_(TUNE) in certain embodiments. The secondary tuning voltage can beapproximately the same as the tuning voltage V_(TUNE) in some otherembodiments.

While the tuning network 47 is illustrated as being a capacitorswitching network, it will be understood that the tuning network canalternatively include any suitable circuit configured to adjust afrequency at which the secondary LC circuit resonates. For example, thetuning network can include any suitable circuit configured toselectively electrically couple one or more passive impedance elementsto the secondary LC circuit. As another example, the tuning network canvary inductance instead of varying capacitance of the secondary LCcircuit. In another example, the tuning network can vary a resistancecoupled to the secondary LC circuit. It will also be understood that thetuning network could include any suitable variable capacitance circuitin place of a capacitor switching network.

The sustaining amplifier of the oscillator 100 illustrated in FIG. 12includes transistors Mn and Mp. Transistors Mn and Mp can implement oneor more features of any of the sustaining amplifiers described herein.While the sustaining amplifier in the oscillator 100 illustrated in FIG.12 includes a differential pair of transistors, it will be understoodthat the principles and advantages described with reference to FIG. 12can be applied to oscillators with other suitable sustaining amplifiers,such as a single ended amplifier.

The bias source of the oscillator 100 illustrated in FIG. 12 includes avariable resistor Rbias. The variable resistor Rbias has a first endcoupled to a power rail (for example, Vdd) and a second end of thesustaining amplifier. As illustrated, the second end of the variableresistor Rbias is coupled to the sources of the transistors Mn and Mp ofthe sustaining amplifier. The resistance of the variable resistor Rbiascan be adjusted to vary an amount of current and/or voltage provided tothe sustaining amplifier. As such, the variable resistor Rbiastransistor can be configured as a current source.

Although n-type or p-type transistors are shown in FIG. 12 forillustrative purposes, it will be understood that the illustratedoscillators can be modified to the mirror image configuration byswapping n-type and p-type transistors, reversing currents, and swappingvoltage rails. In some embodiments, an oscillator can include twosustaining amplifiers, one with n-type devices and one with p-typedevices. One or more of these two sustaining amplifier can includesecondary LC networks that includes any combination of featuresdescribed with reference to the secondary LC network described withreference to FIG. 12.

Dual-LC Oscillator

According to some embodiments, an oscillator can include a primary LCcircuit and a secondary LC circuit. An oscillator with two LC circuitscan be referred to as a dual-LC oscillator. A dual-LC oscillator canoperate in a primary oscillation mode or a secondary oscillation mode,depending on whether oscillation is set by the primary LC circuit or thesecondary LC circuit. Operating the dual-LC oscillator in the primaryoscillation mode can increase the tenability of the oscillator.Operating the dual-LC oscillator in the secondary oscillation mode canreduce the phase noise of the oscillator. Some dual-LC oscillators canoperate in either the primary oscillation mode or the secondaryoscillation mode.

The primary LC circuit can be coupled to the drain of one or moretransistors of a sustaining amplifier and the secondary LC circuit canbe coupled to the gate of the one or more transistors of the sustainingamplifier, for example, as shown in FIGS. 13A and 13B. One or morecoupling capacitors can couple the primary LC circuit to the secondaryLC circuit.

Oscillation of a dual-LC oscillator can be set by selecting capacitancevalues and/or inductance values in the primary circuit LC and/or thesecondary circuit LC. For instance, the primary LC circuit can include afirst tunable capacitance and the secondary LC circuit can also includea second tunable capacitance. Based on adjusting the values of the firsttunable capacitance and/or the second tunable capacitance, oscillationcan be set by either the primary LC circuit or the secondary LC circuit.As another example, the primary LC circuit can include a first tunableinductance and the secondary LC circuit can also include a secondtunable inductance. Based on adjusting the values of the first tunableinductance and/or the second tunable inductance, oscillation can be setby either the primary LC circuit or the secondary LC circuit. Moreover,one of the primary LC circuit or the secondary LC circuit can include atunable capacitance and the other can include a tunable inductance. Insome other implementations, a different tunable passive impedanceelement can alternatively or additionally be included in the primary LCcircuit and/or the secondary LC circuit.

The primary LC circuit can include a first node N1 and a second node N2.The signals at the first node N1 and the second node N2 can besinusoidal signals that are 180 degrees out of phase with respect toeach other, in some implementations. For instance, the first node N1 andthe second node N2 can have voltages that have opposite signs andapproximately the same magnitude at any given time. In someimplementations, the first node N1 and the second node N2 can bereferred to as a non-inverted node and an inverted node, respectively,and the signals at these nodes can be inverted from each other.

A sustaining amplifier can include a first transistor Mn configured todrive the first node N1. For instance, the sustaining amplifier caninclude a first field effect transistor Mn having a drain electricallyconnected to the first node N1. The sustaining amplifier can alsoinclude a second transistor Mp configured to drive the second node N2.For instance, the sustaining amplifier can include a second field effecttransistor having a drain electrically connected to the second node N2.The first transistor Mn and the second transistor Mp of the sustainingamplifier can be a differential pair. The third node N3 can beelectrically connected to a control terminal, such as a gate, of thesecond transistor Mp. The fourth node N4 can be electrically connectedto a control terminal, such as a gate, of the first transistor Mn. Afirst coupling capacitor can be in series between the first node and thethird node. A second coupling capacitor can be in series between thesecond node N2 and the fourth node N4. A bias V_(G) can be a voltage ata power rail voltage (for example, at a ground potential) in someimplementations. The bias V_(G) can be at another suitable voltage levelin some other implementations.

FIGS. 13A and 13B are schematic diagrams of an oscillator 110 withdual-oscillation modes, according to an embodiment. In FIG. 13A, theoscillator 110 is configured to operate in the primary oscillation mode.The oscillator 110 is configured to operate in the secondary oscillationmode in FIG. 13B. The oscillator 110 can be programmable such that itcan operate in either the primary oscillation mode or the secondaryoscillation mode.

Referring to FIG. 13A, the primary oscillation mode of the oscillator110 will be described. In the primary oscillation mode, oscillation isset by the inductance and the capacitance of the primary LC circuit.Accordingly, signals at nodes N1, N2, N3, and N4 can oscillate atfrequency that is primarily based on the inductance and the capacitanceof the primary LC circuit. As illustrated in FIG. 13A, the primary LCcircuit can include a primary capacitor 131 having a primary capacitanceCp and primary inductors 132 and 133 each having a primary inductanceLp. The primary capacitor 131 can represent a tunable capacitive networkthat can be programmable to adjust the primary capacitance. For example,the primary capacitor 131 can represent a switching network 14 and/or atuning network 47 in accordance with the principles and advantagesdescribed herein. The primary capacitor 131 can represent a tunablecapacitive network in parallel with a capacitor, for example, asillustrated by the capacitor 49 and the switching network 14 shown inFIG. 12. Alternatively or additionally, one or more of the primaryinductors 132 or 133 can represent a tunable inductance network that canbe programmable to adjust the primary inductance. In anotherimplementation (not shown), a single differential inductor with acentral tap electrically connected to a power rail, such as a groundpotential or Vdd, can be used in place of the primary inductors 132 and133. As illustrated in FIG. 13A, the secondary LC circuit can include asecondary capacitor 134 having a secondary capacitance Cs and secondaryinductors 135 and 136 each having a secondary inductance Ls. Thesecondary capacitor 134 can represent a tunable capacitive network thatcan be programmable to adjust the secondary capacitance. For example,the secondary capacitance can represent a switching network 14 and/or atuning network 47 in accordance with the principles and advantagesdescribed herein. Alternatively or additionally, one or more of thesecondary inductors 135 or 136 can represent a tunable inductancenetwork that can be programmable to adjust the secondary inductance. Inanother implementation (not shown), a single differential inductor witha central tap electrically connected to a power rail, such as a groundpotential or Vdd, can be used in place of the secondary inductors 135and 136. As shown in FIG. 13A, a first coupling capacitor 137 can be inseries between the first node N1 and the third node N3. A secondcoupling capacitor 138 can also be in series between the second node N2and the fourth node N4. The first coupling capacitor 137 and the secondcoupling capacitor can have a capacitance of Cc.

In the primary oscillation mode, frequency of oscillation is set by theprimary inductance Lp and the primary capacitance Cp. For instance, theinductances Lp and the capacitances Cp can be selected such that theresonant frequency of the primary LC circuit is substantially less thanthe resonant frequency of the secondary LC circuit, which can cause theoscillator 110 to oscillate at about the resonant frequency of theprimary LC circuit. For instance, the secondary capacitance Cs and thesecondary inductances Ls can be selected such that the oscillator shouldnot be able to operate at the resonant frequency of the secondary LCcircuit. In one example, the values of the secondary inductances Ls andthe secondary capacitance Cs can be set such that the secondary LCcircuit has a resonant frequency of about 10 GHz while the values of theprimary inductances Lp and the primary capacitance Cp can be set suchthat the primary LC circuit has a resonant frequency selected in therange from about 3 GHz to 4 GHz.

As shown in FIG. 13A, the amplitude of a voltage at a gate of atransistor in the sustaining amplifier can be less than a voltage at adrain of the transistor in the sustaining amplifier in the primaryoscillation mode. The amplitude of the voltage at a gate of thetransistor in the sustaining amplifier can be attenuated by a factor ofabout

$\frac{1}{1 + \frac{2\; C\; s}{C\; c}},$

in which Cs can represent the secondary capacitance of the secondarycapacitor 134 shown in FIG. 13A and parasitic capacitances of thesustaining amplifier, and Cc can represent the capacitance of thecorresponding coupling capacitor 137 or 138. The oscillator 110 of FIG.13A can output the signal at node N1 and/or the signal at node N2. Theoutput of the oscillator 110 can then be buffered.

Referring to FIG. 13B, the secondary oscillation mode of the oscillator110 will be described. In the secondary oscillation mode, oscillation isset by the inductance and the capacitance of the secondary LC circuit.Accordingly, signals at nodes N1, N2, N3, and N4 can oscillate atfrequency that is primarily based on the inductance and the capacitanceof the secondary LC circuit. In the secondary oscillation mode,oscillation can be set by the inductances Ls of the secondary inductors135 and 136 and the capacitance Cs of the secondary capacitor 134. Forinstance, the inductances Ls and the capacitance Cs in the secondary LCcircuit can be selected such that the resonant frequency of thesecondary LC circuit is substantially less than the resonant frequencyof the primary LC circuit such that the oscillator 110 can oscillate atthe resonant frequency of the secondary LC circuit. In one example, theprimary capacitance Cp of the primary capacitor 131 and the primaryinductances Lp of the primary inductors 132 and 133 can be selected suchthat the oscillator should not be able to operate at the resonantfrequency of the primary LC circuit. For instance, the inductance valuesof the primary inductors 132 and 133 and the capacitance value of theprimary capacitor 134 can be set such that the primary LC circuit has aresonant frequency of about 10 GHz while the inductance values of thesecondary inductors Ls and the capacitance value of the secondarycapacitor Cs can be set such that the secondary LC circuit has aresonant frequency of about 3 GHz to 4 GHz.

As shown in FIG. 13B, the amplitude of a voltage at a gate of atransistor in the sustaining amplifier can be greater than a voltage ata drain of the transistor in the sustaining amplifier in the secondaryoscillation mode. In one embodiment, this can approximately maximize theamplitude of an oscillating signal at the gate of the transistor in thesustaining amplifier. Increasing an amplitude of an oscillating signalat the gate can increase a slope of the oscillating signal, which canthereby reduce phase noise. Accordingly, it can be advantageous tooperate in the second mode of oscillation to reduce phase noise. Theamplitude of the voltage at a drain of the transistor in the sustainingamplifier can be attenuated by a factor of, for example, about

$\frac{1}{1 + \frac{2\; C\; p}{C\; c}},$

in which Cp can represent the primary capacitance of the primarycapacitor 131, and Cc can represent the capacitance of a correspondingcoupling capacitor 137 or 138. Such an attenuation can satisfy fieldeffect transistor breakdown requirements. The oscillator 110 of FIG. 13Acan output the signal at node N3 and/or the signal at node N4. Theoutput of the oscillator 110 can then be buffered.

$\frac{C\; p}{C\; c}$

During secondary oscillation mode of the oscillator 110, the ratio ofcan be approximately maximized in an embodiment. The secondaryoscillation mode can consume more current than the first oscillationmode, in some implementations. To cause the oscillator 110 to operate inthe secondary mode of oscillation, the primary capacitance Cp and/or thesecondary capacitance Cs can be tuned. In one embodiment, the secondarycapacitance Cs can be about 10 times greater than the primarycapacitance Cp to set the oscillator 110 to operate in the secondaryoscillation mode.

In certain embodiments, the oscillator 110 can be tunable such that theoscillator can operate in either the primary oscillation mode or thesecondary oscillation mode and the oscillator can tuned such that ittransitions from operating in one mode to operating in the other mode.According to some other embodiments, the oscillator 110 can beconfigured such that it only operates in one of the primary oscillationmode or the secondary oscillation mode. Moreover, it will be understoodthat an oscillator with a primary LC circuit and a secondary LC circuitcan be implemented with any combination of features discussed herein.

Oscillators with Single-Ended Sustaining Amplifier and Secondary LCCircuit

Although certain features of the oscillator 100 of FIG. 12 and theoscillator 110 of FIGS. 13A and 13B have been described with referenceto a sustaining amplifier that includes a differential pair oftransistors, the principles and advantages described with reference tothese oscillators can be applied to oscillators having a number ofdifferent sustaining amplifier topologies. For instance, one or morefeatures described with reference to FIG. 12, FIG. 13A, and/or FIG. 13Bcan be applied to an oscillator with a single ended sustainingamplifier.

FIG. 14 is a schematic diagram illustrating a tunable passive impedancenetwork configured to bias control an input of a switch in asingle-sided sustaining amplifier, according to an embodiment. Theoscillator 120 illustrated in FIG. 14 has a single-ended sustainingamplifier, instead of a differential sustaining amplifier like in theoscillators 100 and 110. Besides having a single-ended sustainingamplifier and a different passive impedance network, the oscillator 120can be substantially the same and/or functionally similar to theoscillator 100 and/or the oscillator 110 according to certainembodiments.

As illustrated in FIG. 14, the passive impedance network 20 e caninclude a first passive impedance element 122, a second passiveimpedance element 124, and a tunable passive impedance network 126. Afirst end of the first passive impedance element 122 can be coupled toan inverted node (for example, node N2) and a second end of the firstpassive impedance element 122 can be coupled to a control terminal (suchas a gate when Mn in a field effect transistor) of the transistor Mn.The first passive impedance element 122 can be a capacitor according tosome implementations. In the passive impedance network 20 e, the secondpassive impedance element 124 can apply a bias V_(G) to the gate of thetransistor Mn. In some implementations, the second passive impedanceelement 124 can be an inductor. A first end of the second passiveimpedance element 124 can receive the bias V_(G) and a second end of thesecond explicit passive impedance element 32 b can apply the bias V_(G)to the gate of the transistor Mn. The tunable passive impedance network126 can be coupled in parallel with the second passive impedance element124.

The passive impedance elements 122 and 124 can correspond to explicitpassive impedance elements rather than merely parasitic impedances. Thefirst passive impedance element 122 can block a direct current (DC)voltage from being applied to the gate of transistor Mn. The secondpassive impedance element 124 and the tunable passive impedance network126 can set a resonant frequency of an LC circuit coupled between a biasV_(G) and a control terminal, such as a gate, of the transistor Mn. Forinstance, the second passive impedance element 124 and the tunablepassive impedance network 126 can be an LC circuit. The bias V_(G) canbe a voltage at a power rail voltage (for example, at a ground potentialor at Vdd) in some implementations. The bias V_(G) can be at anothersuitable voltage level in some other implementations. The passiveimpedance elements 122 and 124 can include inductors, capacitors,resistors, or other passive circuit elements. The tunable passiveimpedance network 126 can implement any combination of featuresdescribed with reference to the tuning network 47 and/or the switchingnetwork 14, for example. In one embodiment, the first passive impedanceelement 122 comprises a capacitor, the second passive impedance elementcomprises an inductor, and the tunable passive impedance networkcomprises a tunable capacitive network.

CONCLUSION

In the embodiments described above, some methods, systems, and/or weredescribed in conjunction with particular embodiments, such as an LCtank. A skilled artisan will, however, appreciate that the principlesand advantages of the embodiments can be used for any other systems,apparatus, or methods with a need for a low noise oscillator. Someexample systems with a need for a low noise oscillator include wired andwireless communications transceivers, clock and data recovery circuitsfor fiber optic cables, SerDes interfaces, phase-locked loops (PLLs),function generators, frequency synthesizers, and the like.

Such methods, systems, and/or apparatus can be implemented into variouselectronic devices. Examples of the electronic devices can include, butare not limited to, consumer electronic products, parts of the consumerelectronic products, electronic test equipment, etc. Examples of theelectronic devices can also include memory chips, memory modules,circuits of optical networks or other communication networks, and diskdriver circuits. The consumer electronic products can include, but arenot limited to, wireless devices, a mobile phone (for example, a smartphone), cellular base stations, a telephone, a television, a computermonitor, a computer, a hand-held computer, a tablet computer, a personaldigital assistant (PDA), a microwave, a refrigerator, a stereo system, acassette recorder or player, a DVD player, a CD player, a digital videorecorder (DVR), a VCR, an MP3 player, a radio, a camcorder, a camera, adigital camera, a portable memory chip, a washer, a dryer, awasher/dryer, a copier, a facsimile machine, a scanner, a multifunctional peripheral device, a wrist watch, a clock, etc. Further, theelectronic device can include unfinished products.

Unless the context clearly requires otherwise, throughout thedescription and the claims, the words “comprise,” “comprising,”“include,” “including,” and the like are to be construed in an inclusivesense, as opposed to an exclusive or exhaustive sense; that is to say,in the sense of “including, but not limited to.” The words “coupled” orconnected”, as generally used herein, refer to two or more elements thatmay be either directly connected, or connected by way of one or moreintermediate elements. As used herein, “active circuit elements”generally refer to circuit elements that are capable of deliveringenergy, and “passive circuit elements” generally refer to circuitelements that are configured to receive and/or dissipate/store energy.Additionally, the words “herein,” “above,” “below,” and words of similarimport, when used in this application, shall refer to this applicationas a whole and not to any particular portions of this application. Wherethe context permits, words in the Detailed Description using thesingular or plural number may also include the plural or singularnumber, respectively. The words “or” in reference to a list of two ormore items, is intended to cover all of the following interpretations ofthe word: any of the items in the list, all of the items in the list,and any combination of the items in the list.

Moreover, conditional language used herein, such as, among others,“can,” “could,” “might,” “may,” “e.g.,” “for example,” “such as” and thelike, unless specifically stated otherwise, or otherwise understoodwithin the context as used, is generally intended to convey that certainembodiments include, while other embodiments do not include, certainfeatures, elements and/or states. Thus, such conditional language is notgenerally intended to imply that features, elements and/or states are inany way required for one or more embodiments or that one or moreembodiments necessarily include logic for deciding, with or withoutauthor input or prompting, whether these features, elements and/orstates are included or are to be performed in any particular embodiment.

The teachings of the inventions provided herein can be applied to othersystems, not necessarily the systems described above. The elements andacts of the various embodiments described above can be combined toprovide further embodiments.

While certain embodiments of the inventions have been described, theseembodiments have been presented by way of example only, and are notintended to limit the scope of the disclosure. Indeed, the novelmethods, apparatus, and systems described herein may be embodied in avariety of other forms. Furthermore, various omissions, substitutionsand changes in the form of the methods and systems described herein maybe made without departing from the spirit of the disclosure. Theaccompanying claims and their equivalents are intended to cover suchforms or modifications as would fall within the scope and spirit of thedisclosure. Accordingly, the scope of the present inventions is definedonly by reference to the appended claims.

What is claimed is:
 1. An apparatus comprising an oscillator, theoscillator comprising: a primary LC circuit having a first node and asecond node; a sustaining amplifier comprising a first transistorconfigured to drive the first node of the primary LC circuit in responseto an input at a first control terminal of the first transistor; and asecondary LC circuit comprising a tuning network having an adjustableimpedance, the tuning network electrically coupled to the first controlterminal of the first transistor, and the secondary LC circuitconfigured to pass a bias to the first control terminal of the firsttransistor.
 2. The apparatus of claim 1, wherein the oscillator isconfigured to cause oscillation of a signal at the first node to be setby the secondary LC circuit by adjusting the adjustable impedance. 3.The apparatus of claim 2, wherein the primary LC circuit comprises aswitching network configured to selectively adjust an impedance betweenthe first node and the second node, and wherein the oscillator isconfigured to cause the primary LC circuit to set oscillation of thesignal at the first node by adjusting the adjustable impedance of thesecondary LC circuit and adjusting the impedance between the first nodeand the second node with the switching network.
 4. The apparatus ofclaim 3, further comprising a capacitor coupled in parallel with theswitching network.
 5. The apparatus of claim 1, wherein, duringoperation, a signal at the control terminal of the first transistor hasa greater amplitude than the signal at the first node.
 6. The apparatusof claim 1, wherein the tuning network is configured to adjust acapacitance of the secondary LC circuit.
 7. The apparatus of claim 1,wherein tuning network is configured to tune a resonant frequency of thesecondary LC circuit.
 8. The apparatus of claim 1, wherein secondary LCcircuit is configured to cause a tuning range of the oscillator to beextended.
 9. The apparatus of claim 1, wherein the secondary LC circuitis configured to pass the bias to the first control terminal of thefirst transistor via an inductor.
 10. The apparatus of claim 1, whereinthe bias is at a ground potential or a power rail potential.
 11. Theapparatus of claim 1, wherein: the sustaining amplifier furthercomprises a second transistor configured to drive the second node of theprimary LC circuit in response to an input at a second control terminalof the second transistor; and the secondary LC circuit is coupled to thesecond control terminal of the second transistor.
 12. The apparatus ofclaim 11, wherein the tuning network is electrically coupled between thefirst control terminal of the first transistor and the second controlterminal of the second transistor, and wherein the tuning network isconfigured to adjust the impedance between the first control terminal ofthe first transistor and the second control terminal of the secondtransistor.
 13. The apparatus of claim 11, wherein the secondary LCcircuit is configured to cause an amount of time that the firsttransistor drives the first node while the second transistor drives thesecond node to be reduced.
 14. The apparatus of claim 1, wherein theoscillator comprises a voltage-controlled oscillator.
 15. An apparatuscomprising an oscillator, the oscillator comprising: a primary LCcircuit having a first node and a second node; a differential pair offield effect transistors comprising: a first field effect transistorhaving a first drain coupled to the first node and a first gate coupledto the second node via a first capacitor; and a second field effecttransistor having a second drain coupled to the second node and a secondgate coupled to the first node via a second capacitor; and a secondaryLC circuit coupled to the first gate of the first field effecttransistor and the second gate of the second field effect transistor;wherein the oscillator is configured to generate a first signal at thefirst gate having greater amplitude than a second signal at the firstdrain.
 16. The apparatus of claim 15, wherein the secondary LC circuitcomprises a tunable passive impedance network configured to adjust animpedance between the first gate of the first field effect transistorand the second gate of the second field effect transistor.
 17. Theapparatus of claim 16, wherein the primary LC circuit comprises aswitching network configured to selectively electrically couplecapacitance between the first node and the second node.
 18. An apparatuscomprising an oscillator, the oscillator comprising: a sustainingamplifier; a first LC circuit coupled to the sustaining amplifier, thesustaining amplifier configured to sustain oscillation of the first LCcircuit; and a second LC circuit coupled to the sustaining amplifier;wherein the oscillator is configured to operate in at least a firstoscillation mode and a second oscillation mode, wherein oscillation ofthe oscillator is set by the first LC circuit in the first oscillationmode, and wherein oscillation of the oscillator is set by the second LCcircuit in the second oscillation mode.
 19. The apparatus of claim 18,wherein the sustaining amplifier comprises a first transistor configuredto drive a node of the first LC circuit in response to first signalreceived at a control terminal of the first transistor, and wherein thesecond LC circuit is coupled to the control terminal of the firsttransistor.
 20. The apparatus of claim 18, wherein the sustainingamplifier comprises: a first field effect transistor having a firstdrain and a first gate; a second field effect transistor having a seconddrain and a second gate; wherein the primary LC circuit has a first nodeand a second node, wherein the first drain is coupled to the first node,wherein the first gate is coupled to the second node via a firstcapacitor, wherein the second drain is coupled to the second node,wherein the second gate is coupled to the first node via a secondcapacitor, and wherein the second LC circuit is coupled to the firstgate and the second gate.